Switching Power Converter With Efficient Switching Control Signal Period Generation

ABSTRACT

A power control system includes a switching power converter and a controller, and the controller responds to a time-varying voltage source signal by generating a switch control signal having a period that varies in accordance with at least one of the following: (i) the period of the switch control signal trends inversely to estimated power delivered to a load coupled to the switching power converter, (ii) the period of the switch control signal trends inversely to instantaneous voltage levels of the voltage source signal, and (iii) the period of the switch control signal trends directly with a line voltage level of the time-varying voltage source signal. In at least one embodiment, the controller achieves an efficient correlation between the switching period with associated switching losses and the instantaneous power transferred to the switching power converter while providing power factor correction (PFC).

CROSS-REFERENCE TO RELATED APPLICATION

This application claims the benefit under 35 U.S.C. §119(e) and 37 C.F.R. §1.78 of U.S. Provisional Application No. 60/915,547, filed on May 2, 2007 and entitled “Power Factor Correction (PFC) Controller Apparatuses and Methods”.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates in general to the field of electronics, and more specifically to a system and method for voltage conversion using a switching power converter with efficient switching control signal period generation.

2. Description of the Related Art

Many devices utilize electrical power to operate. Power is initially supplied by a power source, such as a public utility company, and power sources generally provide a steady state input voltage. However, the voltage levels utilized by various devices may differ from the steady state input voltage provided by the power source. For example, light emitting diode (LED) based lighting systems, typically operate from voltage levels that differ from voltage level supplied by a public utility company. To accommodate the difference between the voltage from the power source and the voltage utilized by the device, power converters are connected between the power source and the device to convert a supply voltage level from an alternating current (AC) power source to, for example, another AC power source having a voltage level different than the supply voltage level. Power converters can also convert AC power into direct (DC) power and DC power into AC power.

Switching power converters represent one example of a type of power converter. A switching power converter utilizes switching and energy storage technology to convert an input voltage into an output voltage suitable for use by a particular device connected to the switching power converter.

FIG. 1 depicts a power control system 100, which includes a switching power converter 102. Voltage source 101 supplies an AC input “mains” voltage V_(mains) to a full, diode bridge rectifier 103. The voltage source 101 is, for example, a public utility, and the AC mains voltage V_(mains) is, for example, a 60 Hz/120 V mains voltage in the United States of America or a 50 Hz/230 V mains voltage in Europe. The rectifier 103 rectifies the input mains voltage V_(mains). The rectifier 103 rectifies the input mains voltage V_(mains) and supplies a rectified, time-varying, primary supply voltage V_(X) to the switching power converter. The switching power converter 102 provides approximately constant voltage power to load 112 while maintaining a resistive input characteristic to voltage source 101. Providing approximately constant voltage power to load 112 while maintaining an approximately resistive input characteristic to voltage source 101 is referred to as power factor correction (PFC). Thus, a power factor corrected switching power converter 102 is controlled so that an input current i_(L) to the switching power converter 102 varies in approximate proportion to the AC mains voltage V_(mains).

PFC and output voltage controller 114 controls the conductivity of PFC switch 108 so as to provide power factor correction and to regulate the output voltage V_(C) of switching power converter 102. The PFC and output voltage controller 114 attempts to control the inductor current i_(L) so that the average inductor current i_(L) is linearly and directly proportional to the primary supply voltage V_(X). A proportionality constant relates the inductor current i_(L) to the primary supply voltage V_(X), and the proportionality constant is adjusted to regulate the voltage to load 112. The PFC and output voltage controller 114 supplies a pulse width modulated (PWM) switch control signal CS₀ to control the conductivity of switch 108. In at least one embodiment, switch 108 is a field effect transistor (FET), and switch control signal CS₀ is the gate voltage of switch 108. The values of the pulse width and duty cycle of switch control signal CS₀ depend on at least two signals, namely, the primary supply voltage V_(X) and the capacitor voltage/output voltage V_(C). Output voltage V_(C) is also commonly referred to as a “link voltage”. Current control loop 119 provides current i_(RTN) to PFC and output voltage controller 114 to allow PFC and output voltage controller 114 to adjust an average i_(L) current 210 (FIG. 2) to equal a target i_(L) current 208 (FIG. 2).

Capacitor 106 supplies stored energy to load 112 when diode 111 is reverse biased and when the primary supply voltage V_(X) is below the RMS value of the input mains. The value of capacitor 106 is a matter of design choice and, in at least one embodiment, is sufficiently large so as to maintain a substantially constant output voltage V_(C), as established by a PFC and output voltage controller 114. A typical value for capacitor 106, when used with a 400 V output voltage V_(C), is 1 microfarad per watt of maximum output power supplied via switching power converter 102. The output voltage V_(C) remains at a substantially constant target value during constant load conditions with ripple at the frequency of primary supply voltage V_(X). However, as load conditions change, the output voltage V_(C) changes. The PFC and output voltage controller 114 responds to the changes in voltage V_(C) by adjusting the switch control signal CS₀ to return the output voltage V_(C) to the target value. In at least one embodiment, the PFC and output voltage controller 114 includes a small capacitor 115 to filter any high frequency signals from the primary supply voltage V_(X).

The switching power converter 102 incurs switching losses each time switch 108 switches between nonconductive and conductive states due to parasitic impedances. The parasitic impedances include a parasitic capacitance 132 across switch 108. During each period TT of switching switch control signal CS₀, energy is used to, for example, charge parasitic capacitance 132. Thus, switching power converter 102 incurs switching losses during each period TT of switch control signal CS₀.

PFC and output voltage controller 114 controls switching power converter 102 so that a desired amount of power is transferred to capacitor 106. The desired amount of power depends upon the voltage and current requirements of load 112. An input voltage control loop 116 provides a sample of primary supply voltage V_(X) to PFC and output voltage controller 114. PFC and output voltage controller 114 determines a difference between a reference voltage V_(REF), which indicates a target voltage for output voltage V_(C), and the actual output voltage V_(C) sensed from node 122 and received as feedback from voltage loop 118. The PFC and output voltage controller 114 generally utilizes technology, such as proportional integral (PI) compensation control, to respond to differences in the output voltage V_(C) relative to the reference voltage V_(REF). The PFC and output voltage controller 114 processes the differences to smoothly adjust the output voltage V_(C) to avoid causing rapid fluctuations in the output voltage V_(C) in response to small error signals. The PFC and output voltage controller 114 generates a pulse width modulated switch control signal CS₀ that drives switch 108. Prodić, Compensator Design and Stability Assessment for Fast Voltage Loops of Power Factor Correction Rectifiers, IEEE Transactions on Power Electronics, Vol. 12, No. 5, September 1007, pp. 1719-1729 (referred to herein as “Prodić”), describes an example of PFC and output voltage controller 114.

FIGS. 2 and 3 depict respective switching control strategies utilized by typical switching power converters, such as switching power converter 102, to convert the input voltage V_(X) into a power factor corrected output voltage V_(C). FIG. 2 depicts a transition switching strategy, and FIG. 3 depicts a constant period switching strategy. Referring to FIGS. 1 and 2, PFC and output voltage controller 114 controls the conductivity of PFC switch 108. The primary supply voltage V_(X) 202 is, in at least one embodiment, a rectified sine wave. To regulate the amount of power transferred and maintain a power factor close to one, PFC and output voltage controller 114 varies the period TT of switch control signal CS₀ so that the inductor current i_(L) (also referred to as the ‘input current’) tracks changes in primary supply voltage V_(X) and holds the output voltage V_(C) constant. The transition switching strategy 204 illustrates that, as the primary supply voltage V_(X) increases, PFC and output voltage controller 114 increases the period TT of switch control signal CS₀. As the primary supply voltage V_(X) decreases, PFC and output voltage controller 114 decreases the period of switch control signal CS₀. In one embodiment of transition switching strategy 204, the pulse width time T1 is approximately constant.

Time T2 represents the flyback time of inductor 110 that occurs when switch 108 is nonconductive and the diode 111 is conductive. In at least one embodiment, the value of inductor 110 is a matter of design choice. In at least one embodiment, the value of inductor 110 is chosen to store sufficient power transferred from voltage source 101 when switch 108 conducts in order to transfer power to capacitor 106 when switch 108 is non-conductive to maintain a desired output voltage V_(C). For the transition switching strategy 204, the pulse width time T1 plus the flyback time T2 equals the period TT of switch control signal CS₀.

The inductor current i_(L) waveform 206 depicts the general behavior of inductor current i_(L) over time relative to the primary supply voltage V_(X). The inductor current i_(L) ramps ‘up’ during pulse width T1 when the switch 108 conducts, i.e. is “ON”. The inductor current i_(L) ramps down during flyback time T2 when switch 108 is nonconductive, i.e. is “OFF”, and supplies inductor current i_(L) through diode 111 to recharge capacitor 106. Discontinuous conduction mode (DCM) occurs when the inductor current i_(L) reaches 0 during the period TT of switch control signal CS₀. Continuous conduction mode (CCM) occurs when the inductor current i_(L) is greater than 0 during the entire period TT. Transition switching strategy 204 operates switching power converter 102 at the boundary of DCM and CCM by beginning each period of switch control signal CS₀ when the inductor current i_(L) just equals 0. The frequency 1/TT of switch control signal CS₀ is, for example, between 20 kHz and 130 kHz. The period TT of switch control signal CS₀ and, thus, the duration of each cycle of inductor i_(L) depicted in inductor current i_(L) waveform 206 is exaggerated for visual clarity. Transition switching strategy 204 operates the switch 108 at high frequencies when little power is transferred from voltage source 101, such as near the zero crossing 212 of the mains voltage V_(mains) and at light load, i.e. when the power demand of load 112 is light.

The PFC and output voltage controller 114 sets a target current 208 that tracks the primary supply voltage V_(X). When the inductor current i_(L) reaches the target current 208 during the pulse width T1, the switch control signal CS₀ opens switch 108, and inductor current i_(L) decreases to zero during flyback time T2. The average current 210 represents the average inductor current i_(L). The average inductor current i_(L) tracks the primary supply voltage V_(X), thus, providing power factor correction.

Referring to FIG. 3, the constant period switching strategy 302 maintains a constant period TT of switch control signal CS₀ and varies the pulse width T1 of switch control signal CS₀ to control inductor current i_(L). As the primary supply voltage V_(X) increases from 0 to line peak, PFC and output voltage controller 114 decreases the pulse width T1 of switch control signal CS₀. Constant period switching strategy 302 operates switching power converter 102 in DCM so that the flyback time T2 plus the pulse width T1 is less than or equal to the period TT of switch control signal CS₀. Inductor current i_(L) waveform 304 depicts the effects of the constant period switching strategy 302 on the inductor current i_(L) relative to the primary supply voltage V_(X). As with the transition switching strategy 204, for the constant period switching strategy 302, the PFC and output voltage controller 114 sets a target current 208 that tracks the primary supply voltage V_(X). For constant period strategy 302, TT≧(T1+T2), so switching power converter 102 operates in DCM.

PFC and output voltage controller 114 updates the switch control signal CS₀ at a frequency much greater than the frequency of input voltage V_(X). The frequency of input voltage V_(X) is generally 50-60 Hz. The frequency 1/TT of switch control signal CS₀ is, for example, between 10 kHz and 130 kHz. Frequencies at or above 20 kHz avoid audio frequencies and frequencies at or below 130 kHz avoids significant switching inefficiencies.

The constant period switching strategy 302 is not efficient in terms of switching losses versus power delivered to load 112. The transition switching strategy 204 is even less efficient than the constant period switching strategy 302.

SUMMARY OF THE INVENTION

In one embodiment of the present invention, a system includes a controller to generate a switch control signal to control conductivity of a switch included in a switching power converter. Controlling conductivity of the switch causes an input current to the switching power converter to vary in approximate proportion to a time varying voltage source signal supplied to the switching power converter. The controller includes a period generator to determine a period of the switch control signal so that the period of the switch control signal varies in accordance with at least one of:

-   -   (i) the period of the switch control signal trends inversely to         estimated power delivered to a load coupled to the switching         power converter;     -   (ii) the period of the switch control signal trends inversely to         instantaneous voltage levels of the time-varying voltage source         signal; and     -   (iii) the period of the switch control signal trends directly         with a line voltage level of the time-varying voltage source         signal; and         The controller also includes a pulse width generator to         determine a pulse width of the switch control signal in response         to at least one of: (i) the determined period of the switch         control signal, (ii) the instantaneous voltage levels of the         voltage source signal, and (iii) a voltage level of the output         voltage signal of the switching power converter.

In another embodiment of the present invention, a method includes generating a switch control signal to control conductivity of a switch included in a switching power converter. Controlling conductivity of the switch causes an input current to the switching power converter to vary in approximate proportion to a time varying voltage source signal supplied to the switching power converter. The method further includes determining a period of the switch control signal so that the period of the switch control signal varies in accordance with at least one of:

-   -   (i) the period of the switch control signal trends inversely to         estimated power delivered to a load coupled to the switching         power converter;     -   (ii) the period of the switch control signal trends inversely to         instantaneous voltage levels of the voltage source signal; and     -   (iii) the period of the switch control signal trends directly         with a line voltage level of the time-varying voltage source         signal;         The method also includes determining a pulse width of the switch         control signal in response to at least one of: (i) the         determined period of the switch control signal, (ii) a voltage         level of the voltage source signal, and (iii) a voltage level of         the output voltage signal of the switching power converter. The         method further includes providing the switch control signal to         the switching power converter.

In another embodiment of the present invention, an apparatus includes means for generating a switch control signal to control conductivity of a switch included in a switching power converter. Controlling conductivity of the switch causes an input current to the switching power converter to vary in approximate proportion to a time varying voltage source signal supplied to the switching power converter. The apparatus further comprises means for determining a period of the switch control signal so that the period of the switch control signal varies in accordance with at least one of:

-   -   (i) the period of the switch control signal trends inversely to         instantaneous power transferred to the switching power         converter;     -   (ii) the period of the switch control signal trends inversely to         voltage level changes of the voltage source signal; and     -   (iii) the period of the switch control signal trends directly         with a line voltage level of the time-varying voltage source         signal; and         The apparatus also includes means for determining a pulse width         of the switch control signal in response to at least one of: (i)         the determined period of the switch control signal, (ii) a         voltage level of the voltage source signal, and (iii) a voltage         level of the output voltage signal of the switching power         converter.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be better understood, and its numerous objects, features and advantages made apparent to those skilled in the art by referencing the accompanying drawings. The use of the same reference number throughout the several figures designates a like or similar element.

FIG. 1 (labeled prior art) depicts a power control system, which includes a switching power converter.

FIG. 2 (labeled prior art) depicts a transition switching control strategy and the effect of the transition switching control strategy on an inductor current of the switching power converter of FIG. 1.

FIG. 3 (labeled prior art) depicts a constant period switching control strategy and the effect of the constant period switching control strategy on an inductor current of the switching power converter of FIG. 1.

FIG. 4 depicts a power control system having a switching power converter and a control signal period-power transfer correlation strategy module.

FIG. 5 depicts a collection of correlated waveforms that depict a correlation between a primary supply voltage, an inductor current, and transferred power in the power control system of FIG. 4.

FIG. 6 depicts an efficient period-instantaneous primary supply voltage V_(X) correlation strategy.

FIG. 7 depicts correlated waveforms between an inductor current and switch control signal of the power control system of FIG. 4.

FIG. 8 depicts a power factor correction (PFC) and output voltage controller of the power control system of FIG. 4.

FIGS. 9-13 depict efficient period-instantaneous primary supply voltage V_(X) correlation strategies.

FIG. 14 depicts a nonlinear delta-sigma modulator.

FIG. 15 depicts a proportional integrator.

FIGS. 16 and 17 depict respective root mean square value generators.

FIG. 18 depicts another embodiment of a PFC and output voltage controller of the power control system of FIG. 4.

FIGS. 19-21 depict efficient period-power transfer-instantaneous primary supply voltage correlation strategies for multiple primary supply RMS voltages and multiple power transfer percentages.

DETAILED DESCRIPTION

A power control system includes a switching power converter and a controller, and the controller responds to a time-varying voltage source signal by generating a switch control signal having a period that varies in accordance with at least one of: (i) the period of the switch control signal trends inversely to estimated power delivered to a load coupled to the switching power converter, (ii) the period of the switch control signal trends inversely to instantaneous voltage levels of the voltage source signal, and (iii) the period of the switch control signal trends directly with a line voltage level of the time-varying voltage source signal. The power control system also includes a pulse width generator to determine a pulse width of the switch control signal in response to at least one of (i) the determined period of the switch control signal, (ii) the instantaneous voltage levels of the voltage source signal, and (iii) a voltage level of the output voltage signal of the switching power converter. Thus, the period can be determined in accordance with a one-way function, two-way function, or three-way function of the variables: (i) estimated power delivered to a load coupled to the switching power converter, (ii) instantaneous voltage levels of the voltage source signal, and (iii) line voltage level of the time-varying voltage source signal (collectively referred to as the “Period Determination Variables”). A “one-way function” indicates that one of the Period Determination Variables (i), (ii), or (iii) is used to determine the switch control signal period. A “two-way function” indicates that any two of the Period Determination Variables (i), (ii), or (iii) are used to determine the switch control signal period. A “three-way function” indicates that all three of the Period Determination Variables (i), (ii), or (iii) are used to determine the switch control signal period.

For power supplies having a voltage source signal that approximates a sine wave, the switching power converter transfers 80% of the power from the voltage source to the load when a phase angle of the voltage source signal is between 45° and 135°. Switching losses in the switching power converter generally increase as switching periods decrease, or, in other words, switching losses in the switching power converter generally increase as switching frequencies increase. By varying the period of the switch control signal so that the period trends in accordance with the one-way function, two-way function, or three-way function of the Period Determination Variables, in at least one embodiment, the controller achieves an efficient correlation between the switching period with associated switching losses and the Period Determination Variable(s) while providing power factor correction (PFC).

FIG. 4 depicts a power control system 400 having a switching power converter 402 and an efficient control signal period generator 408. In at least one embodiment, switching power converter 402 is configured in the same manner as switching power converter 102. Rectifier 103 rectifies the input voltage V_(IN) supplied by voltage source 404 to generate time varying, primary supply voltage V_(X). In at least one embodiment, voltage source 404 is identical to voltage source 101, and input voltage V_(IN) is identical to the mains voltage V_(mains). Power control system 400 also includes PFC and output voltage controller 406. PFC and output voltage controller 406 generates switch control signal CS₁ using feedback signals representing the primary supply voltage V_(X) and output voltage V_(C). PFC and output voltage controller 406 includes the efficient control signal period generator 408 to efficiently correlate a period TT of switch control signal CS₁ with the Period Determination Variables to, for example, increase the efficiency of power control system 400.

In at least one embodiment, the Period Determination Variables are the: (i) estimated power delivered to load 112, (ii) instantaneous voltage levels of primary supply voltage V_(X), and (iii) line voltage level of primary supply voltage V_(X). In at least one embodiment, the estimated power delivered to load 112 is estimated by multiplying the average output voltage V₀ obtained via voltage control loop 418 and the average output current i_(OUT) of switching power converter 402. In at least one embodiment, the estimated power delivered to load 112 is a value “K” determined by the load power demand estimator 803 of FIG. 8. In at least one embodiment, the instantaneous voltage levels of primary supply voltage V_(X) represent a values of primary supply voltage V_(X) sampled via voltage loop 416 at a rate approximately equal to 1/TT, where 1/TT represents the frequency of switch control signal CS₁. The term “instantaneous” includes delays, such as any transmission and processing delays, in obtaining the sampled value of primary supply voltage V_(X). In at least one embodiment, the line voltage level of primary supply voltage V_(X) represents a measure of the primary supply voltage V_(X) for at least one period of primary supply voltage V_(X). For example, in at least one embodiment, the line voltage level is the root mean square (RMS) of primary supply voltage V_(X), a peak of primary supply RMS voltage V_(X) _(—) _(RMS), or an average of primary supply voltage V_(X). For example, the line voltage in the United States of America is nominally 120 Vrms, and the line voltage in Europe is nominally 230 Vrms, where “Vrms” represents an RMS voltage. In general, the line voltage level and the load power demand will be updated at a rate of 50-240 Hz, and the instantaneous voltage will be updated at the switching frequency of switch 108, i.e. the frequency of switch control signal CS₁.

In at least one embodiment, the efficient control signal period generator 408 includes a control signal period strategy that allows the PFC and output voltage controller 406 to generate a period TT of the switch control signal CS₁ that varies in accordance with at least one of the Period Determination Variables.

FIG. 5 depicts a collection of correlated waveforms 500 that depict a correlation between the primary supply voltage V_(X) 502, the inductor current i_(L) 504, and power 506 transferred from voltage source 404 to switching power converter 402. One-half of the period of primary supply voltage V_(X) occurs between phase angles 0°-45° plus phase angles 135°-180°. The RMS voltage of primary supply voltage V_(X) equals the voltage at phase angles 45° and 135°. Thus, primary supply voltage V_(X) is greater than the primary supply RMS voltage V_(X) _(—) _(RMS) for a time equal to half the period TT of primary supply voltage V_(X) and less than the primary supply RMS voltage V_(X) _(—) _(RMS) for a time equal to half the period of TT. The peak voltage of a sine wave primary supply voltage V_(X) is √2·V_(X) _(—) _(RMS). To provide power factor correction, PFC and output voltage controller 406 generates switch control signal CS₁ so that the average inductor current i_(L) 508 tracks the primary supply voltage V_(X). Power 506 transferred from voltage source 404 to switching power converter 402 equals V_(X)·i_(L). Eighty percent of the power 506 is transferred to switching power converter 402 when primary supply voltage V_(X) is greater than primary supply RMS voltage V_(X) _(—) _(RMS), and twenty percent of the power 506 is transferred when primary supply voltage V_(X) is less than primary supply RMS voltage V_(X) _(—) _(RMS). In other words, 80% of the power 506 is transferred when primary supply voltage V_(X) is between phase angles 45° and 135°, and 20% of the power 506 is transferred in the troughs of primary supply voltage V_(X). In at least one embodiment, the troughs of primary supply voltage V_(X) are below primary supply RMS voltage V_(X) _(—) _(RMS) and, for a sine wave, are between phase angles 0°-45° and between phase angles 135°-180°.

Switching power converter 402 also incurs switching losses each time switch 108 switches between nonconductive and conductive states due to parasitic impedances. During each period TT of switching switch control signal CS₁, power is used to, for example, charge parasitic capacitance 132. Switching power converter 402 incurs switching losses during each period TT of switch control signal CS₁. Thus, the higher the frequency of controls signal CS₁, the higher the switching loss.

Referring to FIGS. 1-5, with respect to the conventional transition switching strategy 204, the frequency of switch control signal CS₀ is highest between phase angles 0°-45° and phase angles 135°-180°. Thus, the conventional transition switching strategy 204 incurs the greatest switching loss during the time of the lowest amount of power transfer from voltage source 101 to switching power converter 102. In at least one embodiment, more than half (>50%) of the switching loss associated with the conventional transition switching strategy 204 occurs during the transfer of 20% of the power from voltage source 101 to switching power converter 102. The constant period switching strategy 302 is somewhat more efficient because only approximately 50% of the switching loss associated with the conventional transition switching strategy 204 occurs during the transfer of 20% of the power from voltage source 101 to switching power converter 102.

In at least one embodiment, the efficient control signal period generator 408 allows the PFC and output voltage controller 406 to improve the efficiency of power control system 400 by increasing the period TT of switch control signal CS₁, or in other words decreasing the switching rate of switch 108, during times of low power transfer to load 112, low instantaneous primary supply voltage V_(X), and/or higher primary supply RMS voltage V_(X) _(—) _(RMS). Table 1 sets forth an exemplary switching loss to power transfer ratio comparison: The actual power savings and optimum switch control signal CS₁ period TT generation strategy depend on power components of power control system 400.

TABLE 1 (45) EXEMPLARY (44) SWITCHING STRATEGY SWITCHING LOSS (46) Transition Switching Strategy 204 (47) >50% switching of switch 108 in the troughs of primary supply voltage V_(X) (48) Constant Period Switching (49) 50% switching of Strategy 302 switch 108 in the troughs of primary supply voltage V_(X) (50) Efficient Control Signal Period (51) <50% switching of Generator 408 switch 108 in the troughs of primary supply voltage V_(X).

As previously stated, in at least one embodiment, the troughs of primary supply voltage V_(X) are below primary supply RMS voltage V_(X) _(—) _(RMS) and, for a sine wave, are between phase angles 0°-45° and between phase angles 135°-180°.

FIG. 6 depicts an exemplary efficient period-instantaneous primary supply voltage V_(X) correlation strategy 600 for efficient control signal period generator 408. Referring to FIGS. 5 and 6, as primary supply voltage V_(X) increases towards a peak voltage √2·V_(X) _(—) _(RMS), the power transfer from voltage source 404 to switching power converter 402 increases nonlinearly. For any given value of primary supply voltage V_(X) and power output by switching power converter 402, there is an optimum switching period TT. The optimum period generally increases in the troughs of primary supply voltage V_(X). If the period TT is too short, there is excess switching loss. If the period TT is too long, there will be excessive loss in resistive parasitics, such as the respective resistances of switch 108 and inductor 110 and in core losses of inductor 110. The efficient period-instantaneous primary supply voltage V_(X) correlation strategy 600 provides a strategy for determining the period TT as a function of the instantaneous primary supply voltage V_(X). The actual value of an optimal value of period TT is a matter of design choice and is, for example, dependent upon the values of the components of switching power converter 402 such as the characteristics of inductor 110, switch 108, capacitor 106, and diode 111 along with the instantaneous primary supply voltage V_(X), the primary supply RMS voltage V_(X) _(—) _(RMS), and the power transferred to load 112. Power control system 400 is, in at least one embodiment, more efficient than conventional power control system 100 because the switching frequency of switch 108 increases as more power is supplied by voltage source 404, thus, the controller achieves an efficient correlation between the switching period with associated switching losses of switch 108.

In at least one embodiment, the switching power converter 402 operates in DCM. The frequency 1/TT of switch control signal CS₁ is, for example, between 10 kHz and 130 kHz. The period TT of switch control signal CS₁ and, thus, the duration of each cycle of inductor i_(L) depicted in inductor current i_(L) waveform 504 is exaggerated for visual clarity.

FIG. 7 depicts exemplary, correlated waveforms 700 between an exemplary inductor current i_(L) and switch control signal CS₁. During the time T1 of each pulse width of switch control signal CS₁, inductor current i_(L) rises as energy is transferred from voltage source 404 to inductor 110. During the flyback time T2, inductor current i_(L) decreases as the inductor stored energy charges capacitor 106. The average inductor current i_(L) _(—) _(AVG) 706 tracks primary supply voltage V_(X) to provide power factor correction.

FIG. 8 depicts a PFC and output voltage controller 800, which represents one embodiment of PFC and output voltage controller 406. PFC and output voltage controller 800 determines switch control signal CS₁ in accordance with the switch control signal generation strategy implemented by control signal period generation strategy module 802. Efficient control signal period generation strategy module 802 represents one embodiment of efficient control signal period generator 408. In at least one embodiment, the control signal period generation strategy module 802 generates TT as a function of at least one of: the instantaneous primary supply voltage V_(X) and the estimated power delivered to load 112. In at least one embodiment, the control signal period generation strategy module 802 generates TT as a function of both the primary supply voltage V_(X) and the estimated power delivered to load 112.

The PFC and output voltage controller 800 determines the period TT and pulse width T1 of switch control signal CS₁ to, for example, provide power transfer efficiency and power factor correction for switching power converter 402. In at least one embodiment, the estimated power delivered to load 112 is represented by “K”, the output value of load power demand estimator 803 in the voltage control loop 418. In at least one embodiment, the square of the pulse width period T1, i.e. T1 ², is determined in accordance with Equation 1:

$\begin{matrix} {{T\; 1^{2}} = {\frac{2 \cdot L}{V_{X\_ RMS}^{2}} \cdot K \cdot {TT} \cdot \left( {1 - \frac{V_{X}}{V_{C}}} \right)}} & 1 \end{matrix}$

“T1” is the pulse width (on-time) of the control signal CS₁. “L” represents an inductor value of inductor 110. V_(X) _(—) _(RMS) represents the primary supply RMS voltage V_(X) _(—) _(RMS). “K” represents an estimate of the power demand of load 112 as determined by load power demand estimator 803. “TT” is the period of control signal CS₁ as generated by control signal period generation strategy module 802. “V_(X)” is a sampled value of the current value of primary supply voltage V_(X). “V_(C)” is a sampled value of the output voltage V_(C). In the preferred embodiment, this calculation will be performed in fixed-point arithmetic with appropriately scaled values and work lengths.

The RMS value generator 804 determines primary supply RMS voltage V_(X) _(—) _(RMS) from a sampled primary supply voltage V_(X). Module 806 receives the primary supply RMS voltage V_(X) _(—) _(RMS) value and determines 2·L/(V_(X) _(—) _(RMS) ²). “2·L/(V_(X) _(—) _(RMS) ²)” represents a scaling factor. Boost factor module 808 determines a boost factor (1−V_(X)/V_(C)). Multiplier 810 multiplies switch control signal CS₁, period TT, the output value of module 806, the output value of boost factor module 808, and estimated power demand K to generate T1 ². Nonlinear delta-sigma modulator 812 determines the pulse width T1 of switch control signal CS₁. Pulse width modulator (PWM) 814 receives the pulse width T1 and period TT and generates switch control signal CS₁ so that switch control signal CS₁ has a pulse width T1 and a period TT.

In at least one embodiment, to ensure that switching power converter 402 operates in DCM, the value L of inductor 110 is set in accordance with Equation [2]:

$\begin{matrix} {L = {V_{\min}^{2}/\left\lbrack {\left( {P_{\max} \cdot J} \right) \cdot \left( {2 \cdot f_{\max}} \right) \cdot {\left\lbrack {1 - {\sqrt{2}\left( \frac{V_{\min}}{V_{cap}} \right)}} \right\rbrack.}} \right.}} & \lbrack 2\rbrack \end{matrix}$

“L” is the value of the inductor 110. “V_(min)” is the minimum expected primary supply RMS voltage V_(X) _(—) _(RMS). “P_(max)” is the maximum power demand of load 112. “J” is an overdesign factor and any value greater than 1 indicates an overdesign. In at least one embodiment, “J” is 1.1. “f_(max)” is a maximum frequency of control signal CS₁. “V_(C)” is a nominal expected output voltage V_(C). The flyback time T2 can be determined in accordance with Equation [3]:

$\begin{matrix} {{T\; 2} = {\frac{V_{X}}{V_{C} - V_{X}}.}} & \lbrack 3\rbrack \end{matrix}$

In at least one embodiment, to avoid saturation of inductor 110, the value L of inductor 110 is chosen so that a peak inductor current, i_(L) _(—) _(PEAK) is greater than or equal to the greatest value of V_(X)·T1/L. Generally, the peak inductor current i_(L) _(—) _(PEAK) occurs at full output power at the peak of primary supply voltage V_(X) during low line voltage operation.

The efficient control signal period generation strategy used by PFC and output voltage controller 406 to determine a period of the switch control signal CS₁ is a matter of design choice and can be set to optimize to the efficiency of switching power converter 402.

Additionally, in at least one embodiment, the range of possible primary supply voltage levels also influences the time of period TT. For example, to remain in DCM operation, the period TT is increased for high line voltage conditions in order to remain in DCM operation.

FIGS. 9-13 depict exemplary efficient period-instantaneous primary supply voltage V_(X) correlation strategies. The particular strategy used to provide an efficient period-instantaneous primary supply voltage V_(X) correlation depends on a number of operational factors such as the component values of a power control system, such as power control system 400, operational frequencies, and power delivered to load 112. FIGS. 9-13 illustrate a variety of strategies that provide efficient period-instantaneous primary supply voltage V_(X) correlation. Other period-instantaneous primary supply voltage V_(X) correlation strategies that inversely relate a trend of the switch control signal CS₁ period and the instantaneous primary supply voltage V_(X) can be used is a matter of design choice based, for example, on operational parameters of a power control system.

FIG. 9 depicts efficient period-instantaneous primary supply voltage V_(X) correlation strategy 900. The period TT decreases linearly from primary supply voltage V_(X) equal to 0 to primary supply voltage V_(X) equal to 0.75·√2·V_(X) _(—) _(RMS) and remains constant until primary supply voltage V_(X) equals √2·V_(X) _(—) _(RMS). The constant period TT above voltage V_(B) sets an upper limit on the switching frequency of switch control signal CS₁ to, for example, prevent excessive switching losses of switch 108.

FIG. 10 depicts efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1000. The efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1000 maintains a constant switch control signal CS_(i) period TT until primary supply RMS voltage V_(X) _(—) _(RMS) equals 0.25·√2·V_(X) _(—) _(RMS), decreases linearly thereafter until primary supply RMS voltage V_(X) _(—) _(RMS) equals 0.75·√2·V_(X) _(—) _(RMS), and then remains constant until primary supply RMS voltage V_(X) _(—) _(RMS) equals √2·V_(X) _(—) _(RMS). The constant period TT above voltage V_(A) sets an upper limit for the switching frequency of switch control signal CS₁ to, for example, prevent excessive switching losses of switch 108. The constant period TT below voltage V_(B) sets a lower limit on the switching frequency of switch 108 to, for example, avoid frequencies in a human audible frequency band.

FIG. 11 depicts efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1100. The efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1100 is a step function, and, thus, period TT need only be determined upon the transition from step to step.

FIG. 12 depicts efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1200. The efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1200 initially increases as primary supply RMS voltage V_(X) _(—) _(RMS) increases from 0 and then nonlinearly decreases as primary supply voltage V_(X) approaches √2·V_(X) _(—) _(RMS). Even though efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1200 briefly increases, efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1200 causes the period TT of the switch control signal CS₁ to trend inversely to the instantaneous primary supply voltage V_(X). In at least one embodiment, the efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1200 causes the inductor 110 to get close to saturation.

FIG. 13 depicts efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1300. The efficient period-instantaneous primary supply voltage V_(X) correlation strategy 1300 generally follows a decreases quadratically until primary supply voltage V_(X) equals √2·V_(X) _(—) _(RMS).

The particular period-power transfer correlation strategy used by efficient control signal period generator 408 is a matter of design choice and can be tailored to meet, for example, efficiency, power factor correction, computation complexity, and component characteristics. In the preferred embodiment, period generator 408 is implemented in digital logic and receives digitized representations of input values. The efficient control signal period generator 408 can generate the switch control signal CS₁ period TT in any of a number of ways. For example, the period-instantaneous primary supply voltage V_(X) strategy used by control signal period generation strategy module 802 can be stored as an algorithm, and control signal period generation strategy module 802 can determine the switch control signal CS₁ period TT in accordance with the algorithm. In another embodiment, the period-power transfer correlation strategy can be stored in an optional memory 816. In at least one embodiment, the memory 816 includes a look-up table that correlates values of the period TT and values of primary supply voltage V_(X). The control signal period generation strategy module 802 can then retrieve the value of period TT based on the value of primary supply voltage V_(X).

In at least one embodiment, the PFC and output voltage controller 800 is implemented as a programmable PFC and output voltage controller as described in U.S. patent application Ser. No. 11/967,275, entitled “Programmable Power Control System”, filing date Dec. 31, 2007, assignee Cirrus Logic, Inc., and inventor John L. Melanson. U.S. patent application Ser. No. 11/967,275 includes exemplary systems and methods and is hereby incorporated by reference in its entirety. As the optimum period depends upon the design choice of switching components, allowing programmability of the efficient period control algorithm allows each design to be optimized for efficiency while utilizing the same integrated circuit embodiment of PFC and output voltage controller 800.

FIG. 14 depicts nonlinear delta-sigma modulator 1400, which represents one embodiment of nonlinear delta-sigma modulator 812. The nonlinear delta-sigma modulator 1400 models a nonlinear power transfer process of switching power converter 402. The nonlinear power transfer process of switching power converter 402 can be modeled as a square function, x². Nonlinear delta-sigma modulator 1400 includes a nonlinear system feedback model 1402 represented by x². The output of feedback model 1402 is the square of delay-by-one quantizer output signal T1, i.e. [T1(n−1]². Delay z⁻¹ 1406 represents a delay-by-one of quantizer output signal T1. Negative [T1(n−1)]² is added to T1 ² by adder 1412. The nonlinear delta-sigma modulator 1400 includes a compensation module 1404 that is separate from quantizer 1408. The nonlinearity compensation module 1404 processes output signal u(n) of the loop filter 1410 with a square root function x^(1/2) to compensate for nonlinearities introduced by the nonlinear feedback model 1402. The output c(n) of compensation module 1404 is quantized by quantizer 1408 to generate pulse width T1 for switch control signal CS₁.

FIG. 15 depicts a proportional integrator (PI) compensator 1500, which represents one embodiment of load power demand estimator 803. The PI compensator 1500 generates the load power demand signal K. The load power demand signal K varies as the difference between a reference voltage V_(REF) and the output voltage V_(C), as represented by error signal e_(v) from error generator 1501, varies. The reference signal V_(REF) is set to a desired value of output voltage V_(C). The PI compensator 1500 includes an integral signal path 1502 and a proportional signal path 1504. The integral signal path 1502 includes an integrator 1506 to integrate the error signal e_(v), and a gain module 1508 to multiply the integral of error signal e_(v) by a gain factor g2 and generate the integrated output signal I_(PW). The proportional path 1504 includes a gain module 1510 to multiply the error signal e_(v) by a gain factor g1 and generate the proportional output signal P_(PW). Adder 1512 adds the integrated output signal I_(PW) and the proportional output signal P_(PW) to generate the load power demand signal K.

The values of gain factors g1 and g2 are a matter of design choice. The gain factors g1 and g2 affect the responsiveness of PFC and output voltage controller 406. Exemplary values of gain factors g1 and g2 are set forth in the emulation code of FIGS. 8-31 of U.S. patent application Ser. No. 11/967,269, entitled “Power Control System Using a Nonlinear Delta-Sigma Modulator with Nonlinear Power Conversion Process Modeling”, filed Dec. 31, 2007, assignee Cirrus Logic, Inc., and inventor John L. Melanson. U.S. patent application Ser. No. 11/967,269 describes exemplary systems and methods and is incorporated herein by reference in its entirety. Faster response times of the PFC and output voltage controller 406 allow the switch control signal CS₁ to more rapidly adjust to minimize the error signal e_(v). If the response is too slow, then the output voltage V_(C) may fail to track changes in power demand of load 112 and, thus, fail to maintain an approximately constant value. If the response is too fast, then the output voltage V_(C) may react to minor, brief fluctuations in the power demand of load 112. Such fast reactions could cause oscillations in PFC and output voltage controller 406, damage or reduce the longevity of components, or both. The particular rate of response by proportional integrator 1500 is a design choice.

FIGS. 16 and 17 depict respective exemplary embodiments of RMS value generator 804. The RMS value of primary supply voltage V_(X) is the square root of the average of the squares of primary supply voltage V_(X). RMS value generator 1600 receives a set {V_(X)} samples of primary supply voltage V_(X) during a cycle of primary supply voltage V_(X) and squaring module 1602 squares each sample of primary supply voltage to determine a set {V_(X) ²}. Low pass filter 1604 determines a mean V_(X) _(—) _(MEAN) ² of the set {V_(X) ²}. Square root module 1606 determines the square root of V_(X) _(—) _(MEAN) ² to determine the primary supply RMS voltage V_(X) _(—) _(RMS).

The RMS value generator 1700 receives the primary supply voltage V_(X) and peak detector 1702 determines a peak value V_(X) _(—) _(PEAK) of primary supply voltage V_(X). Since primary supply voltage V_(X) is a sine wave in at least one embodiment, multiplying V_(X) _(—) _(PEAK) by √2/2 with multiplier 1704 generates primary supply RMS voltage V_(X) _(—) _(RMS). In at least one embodiment, as the exact value of V_(X) _(—) _(PEAK) is not critical, the determination of V_(X) _(—) _(PEAK) by RMS value generator 1700 is generally adequate.

FIG. 18 depicts a PFC and output voltage controller 1800 that represents one embodiment of PFC and output voltage controller 406. In at least one embodiment, multi-way function control signal period generation strategy module 1802 determines the period TT of switch control signal CS₁ as a one-way, two-way, or three-way function of the Period Determination Variables. As primary supply RMS voltage V_(X) _(—) _(RMS) increases the average input current, and hence the average inductor current i_(L) required to supply a given amount of power decreases. For example, for primary supply RMS voltage V_(X) _(—) _(RMS)=120V, to supply 30 watts of power, the input equals 250 mA, i.e. P=V·I. For primary supply RMS voltage V_(X) _(—) _(RMS)=240V, to supply 30 watts of power, the RMS inductor current i_(L) _(—) _(RMS) equals 125 mA. Thus, the period TT of switch control signal CS₁ can be increased with increasing values of primary supply RMS voltage V_(X) _(—) _(RMS), which decreases the frequency of switch control signal CS₁. Decreasing the frequency of switch control signal CS₁ increases the efficiency of power control system 400. In at least one embodiment, PFC and output voltage controller 1800 functions the same way as PFC and output voltage controller 800 except the strategy module 1802 determines the period TT of switch control signal CS₁ as a one-way, two-way, or three-way function of the Period Determination Variables.

FIGS. 19, 20, and 21 depict respective efficient period determination strategies 1900, 2000, and 2100 represents a three-way function of the Period Determination Variables. The three-way function” indicates that all three of the Period Determination Variables are used to determine the period TT of switch control signal CS₁. Referring to FIG. 19, the estimated power delivered to load 112 is greater than half (>50%) of a maximum deliverable power to load 112. As the value of primary supply RMS voltage V_(X) _(—) _(RMS) increases, period determination strategy 1900 increases the value of period TT for a given primary supply RMS voltage V_(X) _(—) _(RMS) value. Additionally, the period TT also trends inversely relative to the instantaneous primary supply voltage V_(X). The period determination strategy 1900 represents one embodiment of an efficient period determination strategy that can be utilized by the V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802. The period-power transfer correlation strategies of FIGS. 10-13 can also be utilized by V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802 by increasing the period TT of switch control signal CS₁ with increasing values of primary supply RMS voltage V_(X) _(—) _(RMS).

FIG. 20 depicts an efficient period determination strategy 2000 that represents a three-way function of the Period Determination Variables. The estimated power delivered to load 112 ranges from greater than 20% to 50% of a maximum deliverable power to load 112. As the value of primary supply RMS voltage V_(X) _(—) _(RMS) increases, period determination strategy 2000 increases the value of period TT for a given primary supply RMS voltage V_(X) _(—) _(RMS) value. Additionally, the period TT also trends inversely relative to the instantaneous primary supply voltage V_(X). The period determination strategy 2000 represents one embodiment of an efficient period determination strategy that can be utilized by the V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802. The period-power transfer correlation strategies of FIGS. 10-13 can also be utilized by V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802 by increasing the period TT of switch control signal CS₁ with increasing values of primary supply RMS voltage V_(X) _(—) _(RMS).

FIG. 21 depicts an efficient period determination strategy 2100 that represents a three-way function of the instantaneous voltage levels of the Period Determination Variables. The estimated power delivered to load 112 ranges from 0% to 20% of a maximum deliverable power to load 112. As the value of primary supply RMS voltage V_(X) _(—) _(RMS) increases, period determination strategy 2000 increases the value of period TT for a given primary supply RMS voltage V_(X) _(—) _(RMS) value. For primary supply RMS voltage V_(X) _(—) _(RMS) equal to 240V, if the relationship between period TT and the instantaneous primary supply voltage V_(X) at √2·240 at a constant rate as primary supply RMS voltage V_(X) _(—) _(RMS) decreased, the period TT would be 80 micro seconds at instantaneous primary supply voltage V_(X) equal 0 V. However, to keep the frequency of switch 108 above 20 kHz, the upper limit of the human audible frequency band, period determination strategy 2100 limits a maximum period TT to 50 micro seconds, i.e. 20 kHz. Additionally, the period TT also trends inversely relative to the instantaneous primary supply voltage V_(X). The period determination strategy 2100 represents one embodiment of an efficient period determination strategy that can be utilized by the V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802. The period-power transfer correlation strategies of FIGS. 10-13 can also be utilized by V_(X) _(—) _(RMS) based efficient control signal period generation strategy module 1802 by increasing the period TT of switch control signal CS₁ with increasing values of primary supply RMS voltage V_(X) _(—) _(RMS).

FIGS. 19-21 taken together depict an exemplary function of the period of the switch control signal switch control signal CS₁ trending inversely to estimated power delivered to load 112. Although a particular embodiment of the estimated power delivered to load 112 and the period TT of switch control signal CS₁ is depicted, the particular relationship where the period TT of switch control signal CS₁ varies inversely to the estimated power delivered to load 112 is a matter of design choice. Additionally, FIGS. 19-21 can be used to as a two-way function of (i) the primary supply voltage V_(X) and (ii) the primary supply RMS voltage V_(X) _(—) _(RMS), while providing power factor correction (PFC) if the estimated power delivered to load 112 is held constant. Additionally, FIGS. 19-21 can be used as a one-way function of the primary supply RMS voltage V_(X) _(—) _(RMS), while providing power factor correction (PFC) by using only inverse relationships between the primary supply RMS voltage V_(X) _(—) _(RMS) and the period TT of switch control signal CS₁.

Thus, PFC and output voltage controller 406 achieves an efficient correlation between the switching period with associated switching losses and (i) the instantaneous power transferred to the switching power converter, (ii) the primary supply voltage V_(X), and/or (iii) the primary supply RMS voltage V_(X) _(—) _(RMS), while providing power factor correction (PFC).

Although the present invention has been described in detail, it should be understood that various changes, substitutions and alterations can be made hereto without departing from the spirit and scope of the invention as defined by the appended claims. 

1. A system comprising: a controller to generate a switch control signal to control conductivity of a switch included in a switching power converter, wherein controlling conductivity of the switch causes an input current to the switching power converter to vary in approximate proportion to a time varying voltage source signal supplied to the switching power converter, wherein the controller comprises: a period generator to determine a period of the switch control signal so that the period of the switch control signal varies in accordance with at least one of: (i) the period of the switch control signal trends inversely to estimated power delivered to a load coupled to the switching power converter; (ii) the period of the switch control signal trends inversely to instantaneous voltage levels of the time-varying voltage source signal; and (iii) the period of the switch control signal trends directly with a line voltage level of the time-varying voltage source signal; and a pulse width generator to determine a pulse width of the switch control signal in response to at least one of: (i) the determined period of the switch control signal, (ii) the instantaneous voltage levels of the voltage source signal, and (iii) a voltage level of the output voltage signal of the switching power converter. 2.-26. (canceled) 